Integrated self-driven active clamp

ABSTRACT

An active clamp circuit includes an active clamp capacitor coupled in series with an active clamp switch and an active clamp controller circuit to receive an active clamp switch current that passes through the active clamp switch and to control the active clamp switch based on the received active clamp switch current. The active clamp controller circuit is configured to enable the active clamp switch based on a first amplitude comparison, the first amplitude comparison being based on the active clamp switch current. The active clamp controller circuit is configured to disable the active clamp switch based on a second amplitude comparison and a third amplitude comparison, the second amplitude comparison and the third amplitude comparison being based on the active clamp switch current.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/913,489, filed on Jun. 26, 2020, which is a continuation of U.S.patent application Ser. No. 16/145,819, filed on Sep. 28, 2018, whichissued as U.S. Pat. No. 10,707,766 on Jul. 7, 2020, and which claimspriority to U.S. Provisional Patent Application No. 62/625,691, filed onFeb. 2, 2018, and entitled “Integrated Adaptive Active Clamp,” all ofwhich are hereby incorporated by reference for all purposes.

BACKGROUND

Switch-mode power supplies (SMPS) are power management components inmodern electronic devices. They provide, among other things, powerefficient and galvanically isolated power to multiple loads. To achievehigh power processing efficiency and/or galvanic isolation,conventionally one or more magnetically coupled elements, semiconductorswitches and associated gate driver circuits are required.

The magnetically coupled elements often suffer from non-trivial leakageinductance phenomena, which necessitate the need for affordable voltagesnubber circuits to control the semiconductor switch peakdrain-to-source voltages. Because of the price-sensitive nature of SMPS,the snubber circuits are conventionally limited to the cost-effectivepassive and power lossy resistor-capacitor-diode (RCD) configurations.

In systems sensitive to power losses and heat generation, thedissipation in lossy components in the form of heat is unsuitable. Thus,recycling of energy using an active clamping configuration within thesystem provides an opportunity for system form-factor reduction andpower efficiency improvement.

Additionally, clamping the maximum drain-source voltages of switchingpower transistors allows for increased device reliability and use ofpower transistors with improved figure-of-merit (FOM). The improved FOMenables the SMPS to operate at higher switching frequency whilemaintaining high power processing efficiency. Furthermore, it allows fora reduction of the SMPS reactive component size and cost.

SUMMARY

In some embodiments, an active clamp circuit includes an active clampcapacitor coupled in series with an active clamp switch and an activeclamp controller circuit. The active clamp controller circuit receivesan active clamp switch current that passes through the active clampswitch and controls the active clamp switch based on the received activeclamp switch current. The active clamp controller circuit is configuredto enable the active clamp switch based on a first amplitude comparison,the first amplitude comparison being based on the active clamp switchcurrent. The active clamp controller circuit is configured to disablethe active clamp switch based on a second amplitude comparison and athird amplitude comparison, the second amplitude comparison and thethird amplitude comparison being based on the active clamp switchcurrent.

In some embodiments, an apparatus includes a power converter circuit toconvert an input voltage from a voltage source to an output voltage. Thepower converter circuit includes a transformer which has a primarywinding and a secondary winding, a first winding node of the primarywinding being coupled to the voltage source to receive the inputvoltage. The apparatus includes a main switch that is coupled to asecond winding node of the primary winding to control a current throughthe primary winding, and an active clamp circuit to clamp a voltage atthe second winding node of the primary winding. The active clamp circuitincludes: i) a series circuit combination of an active clamp capacitorcoupled in series to an active clamp switch, the series circuitcombination being coupled in parallel with the primary winding, and ii)an active clamp controller circuit to receive an active clamp switchcurrent that passes through the active clamp switch and to control theactive clamp switch based on the active clamp switch current. The activeclamp controller circuit is configured to enable the active clamp switchbased on a first amplitude comparison, the first amplitude comparisonbeing based on the active clamp switch current. The active clampcontroller circuit is configured to disable the active clamp switchbased on a second amplitude comparison and a third amplitude comparison,the second amplitude comparison and the third amplitude comparison beingbased on the active clamp switch current.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified circuit schematic of a conventional powerconverter.

FIG. 2 is a simplified circuit schematic of a power converter with aself-driven active clamp circuit, in accordance with some embodiments.

FIG. 3 is a simplified circuit schematic of a self-driven active clampcircuit, in accordance with some embodiments.

FIG. 4 is a simplified plot of signals related to a self-driven activeclamp circuit, in accordance with some embodiments.

FIG. 5A is a simplified plot of signals related to a power converterwith a self-driven active clamp circuit, in accordance with someembodiments.

FIG. 5B is a simplified plot of signals related to a power converterwith a conventional clamp circuit.

FIG. 6 is a simplified circuit schematic of a portion of a powerconverter with a self-driven active clamp circuit, in accordance withsome embodiments.

FIG. 7 is a simplified circuit schematic of a portion of a powerconverter with a self-driven active clamp circuit, in accordance withsome embodiments.

FIG. 8 is a simplified circuit schematic of a portion of a powerconverter with a self-driven active clamp circuit, in accordance withsome embodiments.

FIG. 9A is a simplified plot of signals related to a power converterwith a self-driven active clamp circuit, in accordance with someembodiments.

FIG. 9B is a simplified plot of signals related to a power converterwith a conventional clamp circuit.

FIG. 10 is a screenshot showing experimental results of a powerconverter with a self-driven active clamp circuit, in accordance withsome embodiments.

FIG. 11 is a portion of a process for clamping a voltage of a mainswitch of a power converter using a self-driven active clamp circuit, inaccordance with some embodiments.

FIG. 12 is a portion of a process for clamping a voltage of a mainswitch of a power converter using a self-driven active clamp circuit, inaccordance with some embodiments.

DETAILED DESCRIPTION

Some embodiments described herein provide a self-driven active clampcircuit and self-driven active clamping methods for use in a powerconverter that converts an input voltage to an output voltage using atransformer. In some embodiments, the self-driven active clamp describedherein replaces a diode of a resistor-capacitor-diode (RCD) snubbercircuit of a conventional power converter, advantageously converting theconventional power converter into a power converter having self-drivenactive clamping without needing to change additional control circuits ofthe power converter (such as a primary-side power management integratedcircuit, or a secondary side synchronous switch controller integratedcircuit). In other embodiments, the self-driven active clamp circuit isintegrated into an initial design of a power converter.

The self-driven active clamp circuit advantageously increases powerprocessing efficiency of a power converter by recycling energy stored ina leakage inductance of the transformer. To further improvelight-to-medium power processing efficiency, in some embodiments, theactive clamp circuit is advantageously disabled during light-load andlow-line operation of the power converter (e.g., when a drain-sourcevoltage of a primary side switch is below a voltage threshold, such as500V).

In accordance with some embodiments, the self-driven active clampcircuit clamps a primary side peak voltage of a main switch, whichenables the power converter to utilize primary side and/or secondaryside switches having a lower voltage rating, leading to reduced powerlosses during switch conduction and/or switching. In some embodiments,the self-driven active clamp circuit turns on when a body-diode of anactive clamp field effect transistor (FET) switch of the self-drivenactive clamp circuit begins conducting (e.g., a source-drain current ofthe switch is flowing through the switch) and turns off when near asecond zero crossing of the source-drain current through the switch.Thus, an on-time of the self-driven active clamp circuit can beadvantageously reduced as compared to conventional active clampsolutions because the active clamp switching does not need to coincidewith a main switch turn-on time. As a result, a smaller active clampcapacitor can be utilized as compared to conventional clamping circuitsbecause the active clamp circuit resonant period is shorter. Because theactive clamp capacitor is a high voltage component, using a smallercapacitance can result in significant cost reduction benefits.

At light loads, zero-voltage-switching is conventionally difficult toachieve using conventional active clamp circuits because the periodbetween main switch turn-on times is significantly longer. Embodimentsof the self-driven active clamp circuit described herein allow for nearzero-voltage-switching of the main switch on the primary side of thepower converter across a wide range of output load currents, without theneed for accurate modulation of the transformer magnetizing inductancecurrent value. Such embodiments induce a low-amplitude resonantmagnetizing inductance current, via discharging of the active clampcapacitor, which amplifies the quasi-resonant voltage amplitude. Thisadvantageously results in near zero-voltage switching of the main-switchduring quasi-resonant operation at the first valley, and significantlyreduced main switch turn-on voltage at second and higher valleys.

A further advantage of the self-driven active clamp circuit describedherein is a significantly lower active clamp RMS current (e.g.,reduction of up to 10×), as compared to conventional active clampzero-voltage-switching methods. This enables the self-driven activeclamp circuit to utilize an active clamp switch having a higher RDSON(drain-source on-resistance) as compared to conventional active clampcircuits, resulting in an active clamp circuit which is lower cost,easier to integrate, and having a smaller physical size thanconventional active clamp circuits.

FIG. 1 is a simplified circuit schematic of a conventional powerconverter (“converter”) 100. Some elements of the power converter 100have been omitted from FIG. 1 to simplify the description of powerconverter 100 but are understood to be present. A voltage source V_(in)′is received at the converter 100. V_(in)′ can be provided either as analternating current (AC) or direct current (DC). An input side of theconverter 100 generally includes an input voltage filter block 122, arectifier block 116 (in the case of AC input), an input voltage buffercapacitor C1, an RCD snubber circuit block 114 (which includes acapacitor C2, a resistor R1 and a diode D1), a main switch M1, and amain switch controller circuit (“controller”) 118. The input voltagefilter block 122, rectifier block 116 and the input buffer capacitor C1provide a filtered, buffered, rectified, or otherwise conditioned inputvoltage V_(in) to a transformer 102.

The transformer 102 transfers power from the input side of the converter100 to an output side of the converter 100 and generally includes aprimary winding 104 with a first node 108 and a second node 110, and asecondary winding 106. The output side of the converter 100 generallyincludes an output buffer circuit 112, a synchronous rectifier switchM2, a synchronous rectifier switch controller circuit (“controller”)120, and a load (not shown).

The first node 108 receives V_(in), and the second node 110 is coupledto a terminal of the main switch M1 at the second node 110. The mainswitch M1 controls a current through the primary winding 104 to charge amagnetizing inductance L_(M) of the transformer 102 during a firstportion of a switching cycle of the converter 100. The synchronousrectifier switch M2 controls a current flow through the secondarywinding 106 to discharge the transformer 102 into output buffer circuit112 and/or a load during a subsequent portion of the switching cycle. Insome embodiments, one or both of the main switch M1 and/or thesynchronous rectifier switch M2 are field-effect transistors (FETs),each having a drain node, a source node, and a gate node to control aconduction of current between the drain node and the source node. Inother embodiments, the synchronous rectifier switch M2 is replaced witha diode.

When the main switch M1 is enabled by the controller 118 during thefirst portion of a switching cycle, current flows through the primarywinding 104 to a voltage bias node such as ground. The current flowthrough the primary winding 104 causes energy to be stored in themagnetization inductance L_(M) and a leakage inductance L_(L) of thetransformer 102. When the main switch M1 is disabled in a subsequentportion of the switching cycle, output voltage V_(out) is generated atthe output buffer circuit 112 and is provided to a load (not shown).When the main switch M1 is turned off, a reflected voltage (nV_(out)) isdeveloped at a drain node of the main switch M1 at node 110. Thecontribution of the reflected voltage nV_(out) to a drain-source voltageV_(ds) of the main switch M1 at the node 110 is expressed as:

$\begin{matrix}{V_{ds} = {V_{in} + {nV}_{out}}} & \left( {{Equation}\mspace{20mu} 1} \right)\end{matrix}$

where n is a turns ratio of the transformer 102. Energy stored in theleakage inductance L_(L) of the transformer 102 also contributes to thevoltage V_(ds) developed at the node 110.

The RCD snubber circuit 114 prevents the voltage V_(ds) from increasingto a level that damages the main switch M1. As V_(ds) rises, the diodeD1 becomes forward biased and current flows into the capacitor C2 andinto the resistor R1 to dissipate energy, thereby clamping V_(ds) to alevel that is within a safe operating range of the main switch M1. Tofurther increase the efficiency of the converter 100, the diode D1 canbe replaced with an actively driven clamp switch driven by an activeclamp drive signal. However, conventional active clamping circuitsrequire a control signal or other means of synchronization from thecontroller 118. Thus, a converter 100 that uses a controller 118 that isnot already configured to support conventional active clamping cannoteasily be modified to implement active clamping.

FIG. 2 is a simplified circuit schematic of a power converter(“converter”) 200 with a self-driven active clamp circuit 214, inaccordance with some embodiments. Some elements of the power converter200 have been omitted from FIG. 2 to simplify the description of thepower converter 200 but are understood to be present. The converter 200generally includes the circuit elements discussed with reference toFIG. 1. However, all, or a portion (e.g., the diode D1), of the snubbercircuit 114 of the converter 100 has been replaced with the self-drivenactive clamp circuit (“active clamp circuit”) 214. Advantageously, theactive clamp circuit 214 can replace the snubber circuit 114 of theconverter 100 without making significant modifications to the converter100 (e.g., it does not require a control signal or other synchronizationsignal from the controllers 118, 120). Thus, a converter 100 that wasmanufactured with a snubber circuit similar to the snubber circuit 114can be modified with the self-driven active clamp circuit 214 to performactive clamping. For example, in some embodiments, the diode D1 of theconventional snubber circuit 114 can be replaced with the self-drivenactive clamp circuit 214. A magnetizing inductance current i_(LM)flowing out of the primary winding 104 is described later with referenceto FIG. 5A.

FIG. 3 is a simplified circuit schematic of the self-driven active clampcircuit 214 of the converter 200 introduced with reference to FIG. 2, inaccordance with some embodiments. Some elements of the self-drivenactive clamp circuit 214 have been omitted from FIG. 3 to simplify thedescription of the self-driven active clamp circuit 214 but areunderstood to be present. The active clamp circuit 214 generallyincludes an active clamp switch controller circuit 302 which provides anactive clamp switch control signal to the active clamp switch M3 via agate driver circuit 314. The active clamp circuit 214 additionallyincludes a first current amplitude comparison circuit 304, a secondcurrent amplitude comparison circuit 306, a voltage peak detection andenable circuit 308, a voltage source circuit 316 (e.g., an LDO), anactive clamp switch M3 having a body-diode D2, and an active clampcapacitor C3. The active clamp capacitor C3 is coupled in a series withthe active clamp switch M3, and the series combination of the activeclamp capacitor C3 and the active clamp switch M3 are coupled inparallel with the primary winding 104. The first current amplitudecomparison circuit 304, the second current amplitude comparison circuit306, the voltage peak detection and enable circuit 308, and the activeclamp switch controller circuit 302 are referred to herein as being ofan active clamp controller circuit. The active clamp capacitor C3 canadvantageously be of a lower voltage rating as compared to the voltagerating of capacitor C2 of a conventional RCD circuit 114, thus providingcost savings. For example, in an example embodiment, the active clampcapacitor C3 can have a 250V rating as compared to a 630V rating of thecapacitor C2. In some embodiments, the resistor R1 of the conventionalRCD 114 can advantageously be omitted from the active clamp circuit 214,providing further cost savings.

The voltage source circuit 316 receives a voltage 318 and increases,decreases, or otherwise conditions the voltage 318 to power the gatedriver circuit 314 in order to drive (i.e., enable and disable) theactive clamp switch M3. In some embodiments, the active clamp switch M3is a current-bidirectional two-quadrant switch. In some embodiments, theactive clamp switch M3 is a field-effect transistor (FET) having a drainnode (i.e., a first switch node), a source node (i.e., a second switchnode), and a gate node (i.e., a switch control node) to control aconduction of current between the drain node and the source node. Thedrain node and the source node of the active clamp switch M3 are in aseries circuit combination with the active clamp capacitor C3. The gatenode of the active clamp switch M3 controls a current between the drainnode and the source node of the active clamp switch M3. In someembodiments, the active clamp switch M3 includes a diode, other than abody-diode, which is configured to pass a current between the sourcenode and the drain node (in a first current direction) when the diode isforward biased (e.g., when sufficient voltage is developed across thesource and drain of the active clamp switch M3). When the gate node ofthe active clamp switch M3 is driven by the switch control signal, theactive clamp switch M3 passes current bidirectionally (e.g., in thefirst current direction, and/or a second current direction). In thefirst current direction, current flows from the primary winding 104,through the active clamp switch M3, and into the active clamp capacitorC3. In the second current direction, current flows from the active clampcapacitor C3, through the active clamp switch M3, and into the primarywinding 104.

The voltage peak detection and enable circuit 308 generally includes avoltage comparison circuit 310, a diode D3 and a capacitor C4. Thecapacitor C4 couples a first input of the voltage comparison circuit 310to the node 110 to receive the drain-source voltage Vds of the mainswitch M3. The voltage comparison circuit 310 receives a voltagethreshold 312 (e.g., from a voltage source circuit, not shown) at asecond input. In some embodiments, the voltage threshold 312 is avoltage level above which the main switch M3 could be damaged. In otherembodiments, the voltage threshold 312 is a voltage level thatcorresponds to a heavy load operation of the converter 200. In stillother embodiments, the voltage threshold is a voltage level thatcorresponds to medium or heavy load operation of the converter 200. Thevoltage peak detection and enable circuit 308 transmits an enable signalto the active clamp switch controller circuit 302 when a voltage acrossthe main switch M3 is equal to or surpasses the voltage threshold 312.The voltage peak detection and enable circuit 308 transmits a disablesignal to the active clamp switch controller circuit 302 when a voltageacross the switch M3 does not equal or surpass the voltage threshold312. Thus, the active clamp switch controller circuit is configured todisable the active clamp switch M3 in response to a voltage amplitude ofa voltage developed across the main switch M1 being less than thevoltage threshold 312 (e.g., 500 V), irrespective of a currentsurpassing the first current threshold 320. That is, for as long as theactive clamp switch controller circuit 302 receives a de-asserted signalat the ENABLE input, the active clamp switch M3 is disabled and theoutput PWM signal is not transitioned based on signals received by theactive clamp switch controller circuit 302 at the SET and RESET inputsfrom the current amplitude comparison circuits 304, 306. In contrast,when the voltage comparison signal indicates that the received voltagesurpasses the voltage threshold 312 (ENABLE is asserted), the output PWMsignal is based on the SET and RESET signals received, by the activeclamp switch controller circuit 302, from the current amplitudecomparison circuits 304, 306. Thus, in some embodiments, the self-drivenactive clamp circuit 214 is advantageously disabled during light loadoperation of the converter 200, further increasing power efficiency.

During a portion of the switching cycle when the main switch M1 andactive clamp switch M3 are both off, an active clamp switch currenti_(sd) flows from the primary winding 104, through the body-diode D2, tothe active clamp capacitor C3. During a subsequent portion of theswitching cycle when the main switch M1 is off and the active clampswitch M3 is on, the current i_(sd) oscillates between the active clampcapacitor C3, the magnetizing inductance L_(M), and other intended orparasitic reactive elements of the converter 200.

At a high level, the active clamp switch M3 is enabled if the currenti_(SD) surpasses a first current threshold 320 due to the current i_(SD)flowing from the primary winding 104 to the active clamp capacitor C3through the body diode D2 of the active clamp switch M3 (e.g., bodydiode conduction of the D2 is detected). The active clamp switch M3 isdisabled if a second zero crossing of the current i_(SD) is detected. Afirst zero-crossing of the current i_(SD) is determined to have occurredif the current i_(SD) is less than a second current threshold 322, dueto the current i_(SD) flowing from the active clamp capacitor C3 intothe primary winding 104 through the enabled active clamp switch M3. Thesecond zero-crossing of the current i_(SD) is determined to haveoccurred if the current i_(SD) is greater than the second currentthreshold 322 at a later time, due to current flowing from the primarywinding 104 to the active clamp capacitor C3 through the enabled activeclamp switch M3. Upon determining that the second zero-crossing of thecurrent i_(SD) has occurred, the active clamp switch M3 is disabled.

To elaborate, the first current amplitude comparison circuit 304receives the current i_(sd) at a first input, receives or generates thefirst current threshold 320 at a second input (e.g., from a firstcurrent source circuit, not shown), compares the received current i_(sd)to the first current threshold 320, and transmits a first comparisonsignal to a SET input of the active clamp switch controller circuit 302.In some embodiments, the first current threshold 320 is a currentamplitude that identifies body-diode conduction of the active clampswitch M3. In some embodiments, the first current threshold 320 is about50 mA to 250 mA, such as 100 mA. The second current amplitude comparisoncircuit 306 receives the current i_(sd) at a first input, receives orgenerates the second current threshold 322 at a second input (e.g., froma second current source circuit, not shown), compares the receivedcurrent i_(sd) to the second current threshold 322, and transmits secondand third comparison signals to a RESET input of the active clamp switchcontroller circuit 302. In some embodiments, the second currentthreshold 322 is a current amplitude that identifies a first and secondzero-crossing of the current i_(sd), the RESET input of the active clampswitch controller circuit 302 receiving an asserted second comparisonsignal in response to the second current amplitude circuit 306 detectingthe first zero-crossing of the current i_(sd), and receiving ade-asserted third comparison signal in response to the second currentamplitude circuit 306 detecting the second zero-crossing of the currenti_(sd). In some embodiments, the second current threshold 322 is about−300 mA to 0 mA, such as −50 mA. In some embodiments, the firstzero-crossing of the current i_(sd) is determined by the current i_(sd)transitioning across the second current threshold 322 with a negativegoing slope, and the second zero-crossing of the current i_(sd) isdetermined by the current i_(sd) transitioning across the second currentthreshold 322 with a positive going slope.

Thus, a PWM signal to drive the active clamp switch M3 via the gatedriver circuit 314 is generated by the active clamp switch controllercircuit 302 based on the comparison result signals received at SET andRESET inputs from the comparison circuits 304, 306. The PWM signal istransmitted to the gate driver circuit 314 to control the active clampswitch M3. FIG. 4 provides an example plot 400 illustrating arelationship between the SET and RESET comparison signals 402, 406 and aresultant PWM signal output 404 of the active clamp switch controllercircuit 302, in accordance with some embodiments.

As shown, a rising edge of the SET comparison signal 402 received fromthe first current amplitude comparison circuit 304 (due to detectingbody-diode conduction of M3) triggers a first edge 404 a of the PWMsignal 404. A falling edge of the SET comparison signal 402 does notcause the PWM signal 404 to change state. Likewise, a rising edge of theRESET comparison signal 406 received from the second current amplitudecomparison circuit 306 (due to detecting a first zero-crossing of thecurrent i_(SD)) does not cause the PWM signal 404 to change state.However, a falling edge of the RESET comparison signal 406 (due todetecting a second zero-crossing of the current i_(SD)) triggers asecond edge 404 b of the PWM signal 404. Thus, in the embodiment shown,the active clamp switch M3 is enabled when the current i_(sd) surpassesthe first current threshold 320 (e.g., when the current i_(sd) isflowing through the body diode D2 to the active clamp capacitor C3). Theactive clamp switch M3 is disabled when a second zero-crossing of thecurrent i_(SD) is detected (e.g., when the current i_(sd) is resonatingdue to the magnetizing inductance L_(M), the active clamp capacitor C3,and other reactive components).

In some embodiments, the converter 200 relies on an indirect resonantmechanism to achieve near zero-voltage turn-switching of the main switchM1. FIG. 5A shows a simplified graph 502 which provides operationdetails of the converter 200, in accordance with some embodiments. Incomparison, FIG. 5B shows a simplified graph 550, which providesoperation details of a converter having a conventional active clampcircuit. In both graph 502 and graph 550, the voltage V_(ds) at the node110 and a current flow i_(LM) of the transformer 102 over time t isshown. The current i_(LM) is shown in graph 502 as having an oppositepolarity to that of the current i_(sd) introduced with reference to FIG.3. As was described with reference to FIG. 3, the active clamp switch M3enable signal (PWM) 506 is generated in response to an amplitude of thecurrent flow i_(SD) (negative i_(LM)). The simplified graph 502corresponding to the converter 200 illustrates advantageous zero-voltageswitching of the active clamp switch M3 at region 504 when thebody-diode D2 is conducting, zero-current switching of the active clampswitch M3 at region 508, and near-zero voltage switching of the mainswitch M1 at region 514. The near-zero voltage switching of the mainswitch M1 in region 514 is achieved via a second-order effect arisingfrom the operation of the active clamp circuit 214 as described herein.Specifically, when the active clamp circuit 214 is operational, theactive clamp capacitor C3 voltage is maintained at a voltage valueapproximately equal to the reflected output voltage. As a result, at theinstant when the converter 200 enters region 514 (quasi-resonant mode)the voltage across the active clamp switch M3 is approximately zero andthe equivalent circuit becomes an LC circuit as shown in the simplifiedcircuit schematic 700 of FIG. 7 (L is due to the transformer 102magnetizing inductance and C is the equivalent capacitance Ceq seen atnode 110). Of note is that the equivalent capacitance Ceq as seen atnode 110 is dominated by the respective capacitance of the active clampswitch CossM3 and the active clamp capacitor C3 capacitance (as shown inthe simplified schematic 700 of FIG. 7), which are in the nF range(several orders of magnitude greater than that of a conventionalconverter 100 with the RCD snubber circuit block 114). Due to therelatively large equivalent capacitance Ceq, the resonant magnetizinginductance current amplitude, i_(LM) ¹ (as shown in FIG. 7), is forcedto a larger value (as shown in FIG. 9A) as compared to the case where aconventional RCD snubber is utilized and the voltage at the clampcapacitor is significantly larger than the reflected output voltage (asshown in FIG. 9B). The peak magnetizing inductance current can beapproximated according to the well-known LC resonant conservation ofenergy equation,

L _(m) ·i _(LM1) ² =C _(eq) ·v _(ds) ²   (Equation 2),

where Ceq is the equivalent capacitance at node 110 and i_(LM1) is amagnetizing inductance current (shown in FIG. 7 as i_(LM) ¹). Becausethe amplitude of the magnetizing inductance current is forced to ahigher value initially, and because the equivalent capacitance Ceq asseen at node 110 falls to pF range once the main switch M1 drain-sourcevoltage begins to fall (a simplified schematic 800 of an equivalentcircuit at this point is shown in FIG. 8), the voltage value at thevalleys is forced to a lower voltage value (e.g., 2-3× time less ascompared to a conventional RCD circuit).

Furthermore, a width of the PWM pulse 506 is significantly andadvantageously shorter than a PWM pulse 556 corresponding to aconventional active clamp with zero-voltage switching of the main switchM1 at region 564. An RMS current i_(rms) of the graph 502 issignificantly less than an RMS current i_(rms) of the graph 550, due tothe peak currents being of the same amplitude i_(pk) but the resonantperiod (and duty-ratio D, of the on-time/switching period) of the graph502 being an order of magnitude less than graph 550, as given by theequation,

$\begin{matrix}{{i_{rms}^{2} = {i_{pk}^{2} \cdot \frac{D}{3}}},} & \left( {{Equation}\mspace{20mu} 3} \right)\end{matrix}$

thus, a smaller active clamp capacitor C3 (on the order of nanofarads)can be used, as compared to a conventional active clamp circuit whichmay require an active clamp capacitor being on the order of hundreds ofnanofarads. Based on the near zero voltage crossing at region 514, themain switch M1 is switched on (e.g., resulting in pulse M1 _(EN) 516)when voltage V_(ds) at node 110 is equal to V_(in)−knV_(out), where k isan integer value.

FIGS. 6-8 highlight parasitic capacitive and inductive elements of theconverter 200 which resonate during quasi-resonant operation of theconverter 200, a portion of the switching cycle when the main switch M1and the synchronous rectifier switch M2 are both off. A simplifiedcircuit schematic 600 of FIG. 6 includes elements of the converter 200introduced in FIG. 2, as well as a representation of magnetizinginductance L_(M) 130, parasitic capacitance CossM3 of the active clampswitch M3, parasitic capacitance CossM1 of the main switch M1, andparasitic capacitance CossM2 of the synchronous rectifier switch M2.Also shown is an active clamp switch voltage V_(AC) which is adrain-source voltage of the active clamp switch M3. The dominantcapacitive elements along with the range of their capacitance values fortwo different active clamp switch voltages are illustrated in thesimplified circuit schematics 700, 800 of FIGS. 7-8, respectively.

The simplified circuit schematic 700 of FIG. 7 illustrates particularcapacitive and inductive elements of the converter 200 during a portionof the switching cycle when the voltage across the capacitance CossM3 isapproximately equal to zero volts, the voltage across the active clampcapacitor C3 is approximately equal to nV_(out), and the voltage V_(ds)at node 110 is equal to V_(in)+nV_(out). At this time, the combinedcapacitance of the active clamp capacitor C3 and the parasiticcapacitance CossM3 of the active clamp switch M3 is on the order ofnanofarads. The parasitic capacitance CossM1 of the main switch M1 is onthe order of picofarads, and the parasitic capacitance CossM2 of thesynchronous rectifier switch M2 is on the order of picofarads.

The simplified schematic 800 of FIG. 8 illustrates particular capacitiveand inductive elements of the converter 200 during a portion of theswitching cycle when the voltage across the capacitance CossM3 isapproximately equal to negative nV_(out), and the voltage across theactive clamp capacitor C3 is approximately equal to nV_(out), resultingin the voltage V_(ds) at node 110 being less than the input voltageV_(in). During this portion of the switching cycle, the active clampcapacitor C3 voltage is significantly larger than the reflected outputvoltage nVout. The combined capacitance of the active clamp capacitor C3and the parasitic capacitance CossM3 of the active clamp switch M3 is onthe order of picofarads. The parasitic capacitance CossM1 of the mainswitch M1 is on the order of picofarads, and the parasitic capacitanceCossM2 of the synchronous rectifier switch M2 is on the order ofpicofarads. A current i_(LM) ² of the magnetizing inductance 130 is alsoshown.

The large difference in the capacitive values is important because theinitial large effective capacitance leads to higher magnetizinginductance current (during LC resonance):

$\begin{matrix}{I_{pk} = {V_{peak} \times \sqrt{\frac{C_{eq}}{L_{M}}}}} & \left( {{Equation}\mspace{20mu} 4} \right)\end{matrix}$

where V_(peak) is equal to the reflected voltage nV_(out) on the primaryside of the transformer 102, L_(M) is the magnetizing inductance 130,and Ceq is the equivalent capacitance of the intended and parasiticcapacitances as seen at node 110 shown in FIGS. 6-8. The largedifference in the capacitive values occurs due to a large active clampFET output capacitance CossM3 of the active clamp switch M3 when itsdrain-source voltage is zero, which is only the case when the activeclamp circuit 214 is enabled (e.g., when energy stored in the activeclamp capacitor C3 is recycled during each switching cycle).

A simplified graph 900 of FIG. 9A illustrates voltage V_(ds) at node 110and a magnetizing inductance current i_(LM) (i_(LM) ¹⁻²) of themagnetizing inductance 130 over time t during a quasi-resonant operationof the converter 200, in accordance with some embodiments. A simplifiedgraph 950 of FIG. 9B is a plot of voltage V_(ds) and current i_(LM) overtime t during a quasi-resonant portion of the switching cycle for aconventional lossy RCD snubber circuit. The simplified graph 900illustrates the non-linear effect of Equation 4 when the active clampcapacitor C3 voltage V_(AC) is approximately equal to the reflectedoutput voltage (e.g., V_(AC)˜nV_(out)), as compared to the simplifiedgraph 950 which illustrates a conventional lossy snubber operation wherethe snubber capacitor voltage is significantly greater than thereflected output voltage (e.g., V_(AC)>1.5×nV_(out)).

As shown in the simplified graph 900, the active clamp circuit 214allows for a higher magnetizing current i_(LM) as compared to theconventional lossy snubber circuit shown in the simplified graph 950.The higher magnetizing current i_(LM), in turn, enables more charge tobe removed from the main switch M1, thus leading to lower drain-sourcevoltages V_(ds) during quasi-resonant operation of the converter 200.

FIG. 10 is a screenshot 1000 of experimental results of V_(ds) over timefor a converter 200 having an active clamp circuit 214, in accordancewith some embodiments. A first portion 1002 of the screenshot 1000 showsvoltage V_(ds) at node 110 during operation of the converter 200 whenthe active clamp circuit 214 is disabled. A second portion 1052 of thescreenshot 1000 shows voltage V_(ds) at node 110 during operation of theconverter 200 when the active clamp circuit 214 is enabled and theactive clamp capacitor voltage VAC is approximately equal to thereflected output voltage nVout. As shown, when the active clamp 214 isenabled, the main switch M1 drain-source voltage at the valleys issignificantly smaller (e.g., 100 Vdc vs. 225 Vdc at the first valley;125 Vdc vs. 240 Vdc at the second valley and etc.). The benefit of thereduced valley voltage is due to the reduced main switch CossM1 energydissipated, which is a function of the voltage squared, when the mainswitch M1 is turned on.

FIG. 11 is a portion of an example process for clamping a voltage of amain switch of a power converter (e.g., the power converter 200), inaccordance with some embodiments. The particular steps, order of steps,and combination of steps are shown for illustrative and explanatorypurposes only. Other embodiments can implement different particularsteps, orders of steps, and combinations of steps to achieve similarfunctions or results. At step 1102, an input voltage is received at aprimary winding (104) of a transformer (102) of the power converter(200). At step 1104, a current through the primary winding (104) iscontrolled using a main switch (M1) of the power converter (200). Atstep 1106, a voltage of the main switch (e.g., at node 110) is clampedto a maximum voltage using an active clamp circuit (214). The voltage ofthe main switch (M1) is clamped based on an active clamp switch current(i_(SD)) that passes through an active clamp switch (M3) of the activeclamp circuit (214). Details of step 1106 are presented in FIG. 12, inaccordance with some embodiments. The particular steps, order of steps,and combination of steps are shown for illustrative and explanatorypurposes only. Other embodiments can implement different particularsteps, orders of steps, and combinations of steps to achieve similarfunctions or results.

At step 1202, the active clamp switch current (i_(SD)) is received at afirst current amplitude comparison circuit (304) at a first time. Atstep 1204, it is determined, using the first current amplitudecomparison circuit, if the active clamp switch current is greater than afirst current threshold (320) (e.g., 100 mA). If it is determined atstep 1204 that the active clamp switch current is not greater than thefirst current threshold, flow returns to step 1202. If it is determinedat step 1204 that the active clamp switch current is greater than thefirst current threshold (e.g., a body-diode of the active clamp switchM3 is conducting), flow continues to step 1206. At step 1206, the activeclamp switch (M3) is enabled (e.g., turned on), thereby clamping avoltage at the main switch (M1). At step 1208, the active clamp switchcurrent is received at a second current amplitude comparison circuit(306) at a second time. At step 1210, it is determined, using the secondcurrent amplitude comparison circuit, if the active clamp switch currentis less than a second current threshold (322) (e.g., −50 mA). If it isdetermined at step 1210 that the active clamp switch current is not lessthan the second current threshold, flow returns to step 1208. If it isdetermined at step 1210 that the active clamp switch current is lessthan the second current threshold (e.g., a first zero crossing of theactive clamp switch current), flow continues to step 1212. At step 1212,the active clamp switch current is received at the second currentamplitude comparison circuit at a third time. At step 1214, it isdetermined, using the second current amplitude comparison circuit, ifthe active clamp switch current is greater than the second currentamplitude comparison circuit. If it is determined at step 1214 that theactive clamp switch current is not greater than the second currentthreshold, flow returns to step 1212. If it is determined at step 1214that the active clamp switch current is greater than the second currentthreshold (e.g., a second zero crossing of the active clamp switchcurrent), flow continues to step 1216. At step 1216, the active clampswitch (M3) is disabled (e.g., it is turned off). When the active clampswitch is turned off, the active clamp circuit (214) is no longerclamping a voltage of a main switch (M1) of the power converter (200).

Reference has been made in detail to embodiments of the disclosedinvention, one or more examples of which have been illustrated in theaccompanying figures. Each example has been provided by way ofexplanation of the present technology, not as a limitation of thepresent technology. In fact, while the specification has been describedin detail with respect to specific embodiments of the invention, it willbe appreciated that those skilled in the art, upon attaining anunderstanding of the foregoing, may readily conceive of alterations to,variations of, and equivalents to these embodiments. For instance,features illustrated or described as part of one embodiment may be usedwith another embodiment to yield a still further embodiment. Thus, it isintended that the present subject matter covers all such modificationsand variations within the scope of the appended claims and theirequivalents. These and other modifications and variations to the presentinvention may be practiced by those of ordinary skill in the art,without departing from the scope of the present invention, which is moreparticularly set forth in the appended claims. Furthermore, those ofordinary skill in the art will appreciate that the foregoing descriptionis by way of example only and is not intended to limit the invention.

What is claimed is:
 1. A method, comprising: receiving, at an activeclamp controller circuit, an active clamp switch current that passesthrough an active clamp switch; enabling, using the active clampcontroller circuit, the active clamp switch in response to determining,based on the active clamp switch current, body-diode conduction of theactive clamp switch; and disabling, using the active clamp controllercircuit, the active clamp switch in response to determining, based onthe active clamp switch current, a first zero-crossing of the activeclamp switch current and a second zero-crossing of the active clampswitch current.
 2. The method of claim 1, wherein determining body-diodeconduction of the active clamp switch comprises: comparing, at a firstamplitude comparison circuit of the active clamp controller circuit, theactive clamp switch current to a first current amplitude threshold; andgenerating, using the first amplitude comparison circuit, a firstcomparison signal based on the comparison of the active clamp switchcurrent to the first current amplitude threshold, a level of the firstcomparison signal being indicative of body-diode conduction of theactive clamp switch, the active clamp switch being enabled based on thefirst comparison signal.
 3. The method of claim 2, wherein determiningthe first zero-crossing of the active clamp switch current and thesecond zero-crossing of the active clamp switch current comprises:comparing, at a second amplitude comparison circuit of the active clampcontroller circuit, the active clamp switch current to a second currentamplitude threshold; and generating, using the second amplitudecomparison circuit, a second comparison signal based on the comparisonof the active clamp switch current to the second current amplitudethreshold, a first level of the second comparison signal beingindicative of the first zero-crossing of the active clamp switch and asecond level of the second comparison signal being indicative of thesecond zero-crossing of the active clamp switch, the active clamp switchbeing disabled based on the second comparison signal.
 4. The method ofclaim 3, wherein: determining the first zero-crossing of the activeclamp switch current comprises comparing the active clamp switch currentto the second current amplitude threshold at a first time; anddetermining the second zero-crossing of the active clamp switch currentcomprises comparing the active clamp switch current to the secondcurrent amplitude threshold at a second time.
 5. The method of claim 1,further comprising: clamping, by the active clamp switch, a voltage at anode of another switch to a maximum voltage, the active clamp switchbeing coupled to the node of the other switch.
 6. The method of claim 5,wherein: the node of the other switch is coupled to a winding of atransformer; and the active clamp switch current is a current that flowsbi-directionally, through the active clamp switch, between the windingof the transformer and an active clamp capacitor.
 7. The method of claim1, further comprising: determining, by the active clamp controllercircuit, that a voltage amplitude of a switch voltage developed at anode of another switch is less than a voltage threshold; and disabling,by the active clamp controller circuit, the active clamp switch inresponse to determining that the voltage amplitude of the switch voltageis less than the voltage threshold, irrespective of determiningbody-diode conduction of the active clamp switch.
 8. The method of claim7, further comprising: receiving, at a voltage comparison circuit, theswitch voltage; generating, using the voltage comparison circuit, avoltage comparison signal based on a comparison between the voltageamplitude of the switch voltage and the voltage threshold; disabling, bythe active clamp controller circuit, the active clamp switch in responseto the voltage comparison signal when the voltage comparison signalindicates that the voltage amplitude of the switch voltage does notsurpass the voltage threshold; and not disabling, by the active clampcontroller circuit, the active clamp switch in response to the voltagecomparison signal when the voltage comparison signal indicates that thevoltage amplitude of the switch voltage surpasses the voltage threshold.9. A method, comprising: receiving, at a power converter circuit, aninput voltage from a voltage source; converting, by the power convertercircuit, the input voltage to an output voltage, the power convertercircuit comprising a transformer having a primary winding and asecondary winding, a first winding node of the primary winding beingcoupled to the voltage source to receive the input voltage; controlling,by a main switch of the power converter circuit, a current through theprimary winding, the main switch being coupled to a second winding nodeof the primary winding; receiving, at an active clamp controllercircuit, an active clamp switch current that passes through an activeclamp switch that is coupled to a node of the main switch; enabling,using the active clamp controller circuit, the active clamp switch inresponse to determining, based on the active clamp switch current,body-diode conduction of the active clamp switch; and disabling, usingthe active clamp controller circuit, the active clamp switch in responseto determining, based on the active clamp switch current, a firstzero-crossing of the active clamp switch current and a secondzero-crossing of the active clamp switch current.
 10. The method ofclaim 9, wherein determining body-diode conduction of the active clampswitch comprises: comparing, at a first amplitude comparison circuit ofthe active clamp controller circuit, the active clamp switch current toa first current amplitude threshold; and generating, using the firstamplitude comparison circuit, a first comparison signal based on thecomparison of the active clamp switch current to the first currentamplitude threshold, a level of the first comparison signal beingindicative of body-diode conduction of the active clamp switch, theactive clamp switch being enabled based on the first comparison signal.11. The method of claim 10, wherein determining the first zero-crossingof the active clamp switch current and the second zero-crossing of theactive clamp switch current comprises: comparing, at a second amplitudecomparison circuit of the active clamp controller circuit, the activeclamp switch current to a second current amplitude threshold; andgenerating, using the second amplitude comparison circuit, a secondcomparison signal based on the comparison of the active clamp switchcurrent to the second current amplitude threshold, a first level of thesecond comparison signal being indicative of the first zero-crossing ofthe active clamp switch, a second level of the second comparison signalbeing indicative of the second zero-crossing of the active clamp switch,the active clamp switch being disabled based on the second comparisonsignal.
 12. The method of claim 11, wherein: determining the firstzero-crossing of the active clamp switch current comprises comparing theactive clamp switch current to the second current amplitude threshold ata first time; and determining the second zero-crossing of the activeclamp switch current comprises comparing the active clamp switch currentto the second current amplitude threshold at a second time.
 13. Themethod of claim 9, further comprising: clamping, by the active clampswitch, a voltage at the node of the main switch to a maximum voltage.14. The method of claim 9, wherein: the active clamp switch current is acurrent that flows bi-directionally through the active clamp switchbetween the primary winding of the transformer and an active clampcapacitor.
 15. The method of claim 9, further comprising: disabling,using the active clamp controller circuit, the active clamp switch inresponse to determining that a voltage amplitude of a switch voltagedeveloped at the node of the main switch is less than a voltagethreshold, irrespective of determining body-diode conduction of theactive clamp switch.
 16. The method of claim 15, further comprising:receiving, by a voltage comparison circuit, the switch voltage;outputting, by the voltage comparison circuit, a voltage comparisonsignal based on a comparison between the voltage amplitude of the switchvoltage and the voltage threshold; disabling the active clamp switch inresponse to the voltage comparison signal when the voltage comparisonsignal indicates that the voltage amplitude of the switch voltage doesnot surpass the voltage threshold; and not disabling the active clampswitch in response to the voltage comparison signal when the voltagecomparison signal indicates that the voltage amplitude of the switchvoltage surpasses the voltage threshold.
 17. An apparatus, comprising:an active clamp switch; and an active clamp controller circuit coupledto the active clamp switch and being operable to receive an active clampswitch current that passes through the active clamp switch; wherein: theactive clamp controller circuit is operable to enable the active clampswitch in response to determining, based on the active clamp switchcurrent, body-diode conduction of the active clamp switch; and theactive clamp controller circuit is operable to disable the active clampswitch in response to determining, based on the active clamp switchcurrent, a first zero-crossing of the active clamp switch current and asecond zero-crossing of the active clamp switch current.
 18. Theapparatus of claim 17, further comprising: a first amplitude comparisoncircuit of the active clamp controller circuit; wherein: the firstamplitude comparison circuit is operable to compare the active clampswitch current to a first current amplitude threshold; and the activeclamp controller circuit is operable to generate a first comparisonsignal based on the comparison of the active clamp switch current to thefirst current amplitude threshold, a level of the first comparisonsignal being indicative of body-diode conduction of the active clampswitch, the active clamp switch being enabled based on the firstcomparison signal.
 19. The apparatus of claim 18, further comprising: asecond amplitude comparison circuit of the active clamp controllercircuit; wherein: the second amplitude comparison circuit is operable tocompare the active clamp switch current to a second current amplitudethreshold; and the active clamp controller circuit is operable togenerate a second comparison signal based on the comparison of theactive clamp switch current to the second current amplitude threshold, afirst level of the second comparison signal being indicative of thefirst zero-crossing of the active clamp switch, a second level of thesecond comparison signal being indicative of the second zero-crossing ofthe active clamp switch, the active clamp switch being disabled based onthe second comparison signal.
 20. The apparatus of claim 17, furthercomprising: a third amplitude comparison circuit of the active clampcontroller circuit; wherein: the third amplitude comparison circuit isoperable to compare a voltage amplitude of a switch voltage at a node ofanother switch to a voltage threshold; and the active clamp controllercircuit is operable to disable the active clamp switch in responsedetermining that the voltage amplitude of the switch voltage is lessthan the voltage threshold, irrespective of determining body-diodeconduction of the active clamp switch.